Receiver arrangement with AC coupling

ABSTRACT

A receiver arrangement with AC coupling is specified in which a filter arrangement ( 3 ) is provided in a baseband signal processing chain in a homodyne receiver and can be switched between at least two high-pass filter cut-off frequencies. In this case, a brief changeover is made to a higher cut-off frequency when varying the gain of a low-noise baseband amplifier ( 2 ), for example when the received field strength changes, during the reception mode. The described arrangement allows changes to be carried out to the gain in baseband during the normal reception mode. The present receiver is accordingly suitable for code division multiple access methods, such as those which are provided in the UMTS Standard.

REFERENCE TO RELATED APPLICATIONS

This application is a Continuation of U.S. patent application Ser. No.10/481,535 filed Dec. 19, 2003, entitled “RECEIVER ARRANGEMENT WITH ACCOUPLING,” which claims priority to PCT Application Serial No.PCT/DE02/02266 filed Jun. 21, 2002, entitled “RECEIVER ARRANGEMENT WITHAC COUPLING,” which claims priority to the German Application Serial No.DE 10131676.3 filed Jun. 29, 2001.

FIELD OF INVENTION

The present invention relates to a receiver arrangement with ACcoupling.

BACKGROUND

A distinction is drawn between homodyne and heterodyne architectures formobile radio receivers. While, in the case of heterodyne mobile radioreceivers, a received radio-frequency signal is first of all convertedto an intermediate frequency in order subsequently to be converted tobaseband, homodyne mobile radio receivers convert the radio-frequencysignal to baseband in only one conversion step. Homodyne mobile radioreceivers such as these are also referred to as zero-IF or as directconversion (DC) receivers and are used, for example, in the so-calledthird-generation mobile radio standard, Universal MobileTelecommunications System, UMTS.

One system-dependent disadvantage of direct conversion is DC voltageoffsets which, on the one hand, may be of a static nature, and on theother hand may be of a dynamic nature. The static offsets are causedinter alia by circuitry-dependent offsets in the individual blocks ofthe receiver chains, for example as a result of large pairing tolerancesof the components.

If the received signal is weak, that is to say the described DC offsetsmay be many times higher than the actual useful signal, the offsets arealso amplified with the baseband amplification that is required for theuseful signal, so that a signal which is too large for digitizationwould be produced at the output of the analog baseband chain and at theinput of the analog/digital converter that is normally provided there.It is therefore essential to use circuitry measures to compensate forsuch DC voltage offsets. Conventional methods to compensate for a DCvoltage offset in the analog baseband chain are based either on thehigh-pass filter principle and are provided by means of simple ACcouplings, or feedback loops are provided, with the feedback path havinglow-pass filter characteristics.

Overall, these methods have the disadvantage that, on the one hand, alow cut-off frequency is required for the high-pass filter in order toavoid excessively distorting the useful signal, while, on the otherhand, a high cut-off frequency is required in order to ensure that thestabilization time of the AC coupling is not too long. Furthermore, manydirect converters have adaptive gain control which is dependent on thereceived field strength of the radio-frequency signal. However, gaincontrol systems such as these result in sudden changes in the gain,resulting in transient equalization processes which can exceed theuseful signal by many times, so that the analog/digital converter cannotbe driven ideally.

The described problems relating to transient equalization processesresulting from changes to gain factors are exacerbated by the fact thatsuch transients would be amplified many times further by subsequentamplifier stages, for example programmable amplifiers.

In the case of mobile radio methods such as GSM, which operate usingTime-Division Multiple access TDMA and accordingly transmit and receivein time slots, the described problems can be avoided by making changesto the gain only between time slots. In the case of mobile radio methodsfor which continuous reception is required, for example in the case ofsystems which operate using CDMA, Code Division Multiple Access methods,it is however, desirable to match the gain to the received fieldstrength even when the receiver is being operated without any pauses.

The object of the present invention is to specify a receiver arrangementwith AC coupling in which it is possible to match the gain in thereceiver during a normal reception mode.

SUMMARY

According to the invention, the object is achieved by a receiverarrangement with AC coupling, comprising

an input for supplying a radio-frequency signal,

a frequency converter which is coupled to the input and produces abaseband signal at its output,

a baseband amplifier with variable gain, which is connected to theoutput of the frequency converter and has a control input for varyingthe gain,

a filter arrangement for AC coupling, with an input which is connectedto an output of the baseband amplifier and with a high-pass filter forfiltering the baseband signal, with a cut-off frequency which can beswitched between at least two values, with the lower of the at least twocut-off frequencies being used in a normal mode and with the higher ofthe at least two cut-off frequencies being used in a changeover mode,and with a control input for varying the cut-off frequency, and

a control circuit for activation of the higher cut-off frequency duringthe changeover mode on the basis of a readjustment of the gain of thebaseband amplifier, with an output which is connected to the controlinputs of the baseband amplifier and of the filter arrangement.

The high-pass filter which is provided for AC coupling in the basebandsection of the signal processing chain allows the received useful signalto be transmitted with a high degree of accuracy with a lower cut-offfrequency which is set during normal operation, for example of 2 KHz. Ifthe gain ratio of the amplifier is changed, for example if the receivedfield strength changes, the process is in contrast switched to a highercut-off frequency of, for example, >1 MHz, thus allowing the filter tostabilize more quickly. The higher cut-off frequency is in this caseactivated for a time interval which can be defined, for example of a fewmicroseconds, and the process then switches back again to the lowercut-off frequency, that is to a longer time constant.

The described capability to switch the cut-off frequency for the ACcoupling thus allows matching or slaving of the amplifier power to thereception conditions for example of a mobile station, so that thereceiver present is suitable for mobile radio methods which do notoperate with time slots but in which continuous reception operation mustbe ensured.

The described change to the gain of the baseband amplifier, which ispreferably designed as a low-noise amplifier, with variable gain ispreferably carried out using so-called soft switching, that is to saywithout any abrupt, sudden transition.

Since the described arrangement makes it possible to drive ananalog/digital converter that is connected downstream from the receiverarrangement at an optimum operating point, this analog/digital convertercan be constructed to be less complex and with less resolution.

The described arrangement allows changes to the gain during normalreception.

In one preferred embodiment of the present invention, the controlcircuit is designed such that the higher cut-off frequency is activatedfor a variable time interval, which starts at the time of the change tothe gain of the baseband amplifier. After a change to the baseband gain,the process is switched to the higher cut-off frequency for a fewmicroseconds, for example, in order to allow rapid stabilization and atransient equalization process of short duration.

In a further preferred embodiment of the invention, the higher cut-offfrequency is greater than or equal to 1 MHz.

In a further preferred embodiment of the present invention, the filterarrangement comprises a low-pass filter which can be connected and iscoupled to the control circuit in order to activate the low-pass filtereffect during the changeover mode.

The low-pass filter which can be connected and is preferably connectedupstream of the high-pass filter avoids a sampling effect when thehigher cut-off frequency is activated, as can occur in particular whenthe useful signal is not significantly less than the sudden DC voltagechange that is caused by amplifier switching. A baseband signal that hasbeen subjected to low-pass filtering is in this case supplied to thehigh-pass filter with the higher cut-off frequency during the activationof the higher cut-off frequency. This low-pass filtering effect reducesthe sampling effect in particular in the case of large adjacentchannels, that is to say when the filter arrangement is connectedupstream of the channel filter in baseband. However, the described ACcoupling with the filter arrangement is in fact required at the start ofthe baseband chain in order to eliminate DC offsets which occur as aresult of the frequency conversion and are caused, for example, bymismatches in the mixer.

programmable In a further preferred embodiment of the present invention,a programmable amplifier is provided, is coupled to the output of thefilter and is connected to the control circuit, in order to set thelowest gain which can be set during the changeover mode.

With the described activation of the smallest gain which can be set forthe amplifier, the suppression of a transient that is caused byswitching of the gain factor can be reduced further. The duration ofsuch blanking of the programmable amplifier in this case depends on thestabilization time of a channel filter which is provided in the basebandsection, and preferably corresponds to the time duration of theactivation of the higher cut-off frequency in the filter arrangement bymeans of the control circuit. By way of example, this may last for a fewmicroseconds.

As an alternative to the described blanking of the programmableamplifier, further means for AC coupling may be inserted into thebaseband chain. The described setting of the smallest gain which can beset for the programmable amplifier avoids any additional complexity,however, since the programmable amplifier is normally provided in anycase and invariably has a control input for setting or programming thegain ratio.

In a further preferred embodiment of the present invention, a basebandfilter is provided, and is connected between the filter arrangement andthe programmable amplifier. The baseband filter is used in particularfor channel filtering, that is to say for suppression of undesirableadjacent channels.

In a further preferred embodiment of the present invention, the receiverarrangement is designed for processing balanced signals.

If a local oscillator signal for down-mixing the radio frequency to thebaseband frequency supplied to the described frequency converter is onthe one hand left unchanged and is on the other hand supplied to thedescribed frequency converter with a phase shift of 90.degree., and ifthe baseband chain is thus designed for transmitting complex-valuesignals, then both the in-phase signal path and the quadrature signalpath can each be designed as balanced signal paths. In this case, thedescribed baseband chain, with a frequency converter, a low-noisebaseband amplifier with an adjustable gain ratio, a filter arrangement,a baseband channel filter and a programmable amplifier, is provided inboth the I signal branch and in the Q signal branch.

In a further preferred embodiment of the present invention, thehigh-pass filter in the filter arrangement comprises a parallel branchin a balanced signal path, which has a resistor with aparallel-connected first switch. Furthermore, a series branch is in eachcase connected upstream of the parallel branch in the two lines whichare designed for the transmission of balanced signals, and each have acapacitance.

The high-pass filter, which is constructed in analog form with resistorsand capacitances, can thus be switched in a simple manner by means ofthe described first switch between two time constants and thus betweentwo lower cut-off frequencies for the high-pass filtering. Instead ofthat described, other analog circuits may also be provided in order toform a high-pass filter.

The first switch is in this case opened during normal operation, so thatthe resistance in the parallel branch is effective, accordinglyactivating a long time constant of, for example 2 KHz. In the changeovermode, the first switch is closed, thus resulting in a high cut-offfrequency of, for example, 1 MHz.

In a further preferred embodiment of the present invention, the seriesbranches of the balanced signal path each comprise a parallel circuitformed by a resistor and a second switch, and these are in each caseconnected upstream of the described capacitances in the series branches.

The second switches are closed during normal operation, so that theseries resistances are bridged by a short circuit. In a changeover mode,on the other hand, the second switches are opened, so that thecapacitance C across the series resistance follows signal changes at theinput, thus achieving low-pass filtering. This low-pass filtering avoidsany sampling effect that may occur. Instead of that described, someother analogously constructed low-pass filtering may also be provided,and is connected upstream of the high-pass filter with the switchablecut-off frequency.

In a further preferred embodiment of the present invention, the controlcircuit is designed such that in order to activate the changeover mode,the second switches are first of all opened and the first switches arethen closed, and in that, in order to return to the normal mode, thefirst switches are first of all opened, and the second switches are thenclosed.

The described switching sequence is necessary in order to reliably avoidthe sampling effect. Accordingly, the low-pass filtering is activatedbefore the high-pass filter is switched to the higher cut-off frequencyand, when switching back to normal operation, is first of all switchedback to the low cut-off frequency for high-pass filtering, only afterwhich is the low-pass filtering cancelled.

Further details of the invention are the subject matter of the dependentclaims.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be explained in more detail in the following textusing a number of exemplary embodiments and with reference to thedrawings, in which:

FIG. 1 shows a first exemplary embodiment of the high-pass filter forthe filter arrangement.

FIG. 2 shows the reduction in the equalization processes which can beachieved with the switchable cut-off frequency of the arrangement shownin FIG. 1 when changing the gain ratio.

FIG. 3 shows the transient response of a baseband signal processingchain with a filter arrangement as shown in FIG. 1, as well as adownstream channel filter and a programmable amplifier on the basis of astep-function response at the output of the programmable amplifier,

FIG. 4 uses a signal profile at the baseband output to show the samplingeffect which occurs in particular when the useful signal is notsignificantly smaller than the sudden DC change which occurs byswitching the gain at the start of the baseband chain.

FIG. 5 uses an example to show a development of the filter arrangementshown in FIG. 1 with additional low-pass filtering in order to avoid asampling effect,

FIG. 6 shows the signal profile as shown in FIG. 5 by analogy to FIG. 4,

FIG. 7 uses a block diagram to show the receiver arrangement with afilter arrangement for example as shown in FIG. 1 or 5, and

FIG. 8 shows the signal profiles of the control signals of the basebandamplifier, filter arrangement and programmable amplifier as shown in theblock diagram in FIG. 7.

DETAILED DESCRIPTION

FIG. 1 shows a filter arrangement for AC coupling for use in a balancedreceiver, to be precise in its baseband signal processing chain andpreferably downstream from a low-noise variable-gain baseband amplifierwhich is arranged at the signal output of a frequency converter fordirect conversion.

The arrangement shown in FIG. 1 has a respective series capacitance C1,C2 in the two series branches of the balanced baseband signal path. Onthe output side a respective resistor R1, R2 is connected to thecapacitances, C1, C2, with the resistors R1, R2 which are connected inseries with one another and form a parallel branch in the basebandsignal path, being directly connected to one another by a furtherconnection in each case. A first respective switch S1, S2 is connectedin parallel with the first resistors R1, R2.

The arrangement shown in FIG. 1 is a high-pass filter with a switchablecut-off frequency, with the cut-off frequencies being formed from thecomponent values of the capacitances C1, C2, of the resistors R1, R2 andfrom parasitic elements. The switches S1, S2 are preferably switched atthe same time and are connected to a control circuit.

The circuit shown in FIG. 1 is preferably suitable for use in receiverarrangements which are based on the direct conversion principle, that isto say for homodyne reception architectures.

The circuit shown in FIG. 1 in this case allows continuous receptionoperation even when the gain factor of an amplifier which is connectedupstream of the AC coupling is changed during normal reception operationfor matching the received field strength. For this purpose, a changeoveris made to a considerably higher cut-off frequency when the gain factoris changed, for example by closing the switches S1, S2, so that a suddenDC change which is caused by the switching of the gain decays with avery short time constant. The cut-off frequency of the high-pass filterthat is produced when the switches S1, S2 are open may in this case bechosen to be very low, for example around 2 KHz, so that the usefulsignal is transmitted virtually without any distortion. Overall, only aminimum amount of the information is lost from the useful signal when asudden change in gain occurs. Any transients which may occur whenswitching the gain remain negligibly small. An analog/digital converterwhich is connected downstream from a baseband chain in a receiver withAC coupling as shown in FIG. 1 can thus be driven at the optimumoperating point, so that the converter can be designed to have aparticularly small number of bits for digitization. Since sudden gainchanges can be carried out during normal reception, the described ACcoupling is suitable for code division multiple access methods, asenvisaged in the UMTS Standard.

FIG. 2 shows the time-domain profile of the voltage in microvoltsplotted against the time in microseconds for a signal A at the output ofthe filter arrangement shown in FIG. 1, when an amplifier to whose gaina change is made is arranged at its input. The filter arrangement shownin FIG. 1 in this case has a low cut-off frequency of 2 KHz, and a highcut-off frequency of >1 MHz. The DC offset is in this case about 5 mV,and the gain changes suddenly from 18 dB to 12 dB, with the −6 dB suddenchange in the upstream amplifier being made with a soft transition. Ascan be seen, the stabilization time for the filter arrangement has beenreduced to <1 microsecond, and the transient relating to the switchingtime t=1 .mu.s is restricted to 0.6 mV. Without any switching of thecut-off frequency for a filter with a fixed cut-off frequency of 4 KHz,a DC offset of 5 mV and a sudden voltage gain change likewise from 18 to12 dB, the high-pass filter would stabilize only very slowly, with thestabilization time being more than 100 .mu.s. Furthermore, a transientof 20 mV would occur which would still be amplified many times, overallup to 50 dB, by downstream amplifier stages.

FIG. 3 shows the step-function response B of a receiver arrangement witha low-noise variable-gain baseband amplifier and the downstream filterarrangement as shown in FIG. 1, measured at the output of a programmableamplifier which is connected downstream of a channel filter which is inturn connected to the output of the filter arrangement. This shows thatthe stabilization time of this overall baseband chain when a change ismade to the gain factor of the baseband amplifier at the input of thebaseband chain is restricted to about 3 microseconds and that themaximum transient that occurs in the process and which does not exceed 4mV can, as an approximation, be ignored.

FIG. 4 shows the signal relationships on the basis of a signal profile Cin the situation where the actual useful signal is not considerably lessthan the sudden DC change that is caused by switching the gain. Duringthe changeover mode, that is to say when the switches S1, S2 are closedas shown in FIG. 1 and in which the high-pass filter has the highcut-off frequency, the voltage across the capacitors C1, C2 follows theinput signal. When switching back to normal operation by opening theswitches S1, S2, a sampling effect occurs in this case, that is to saythe instantaneous signal across the capacitors C1, C2 is sampled anddecays with the long time constant, that is now selected once again, fornormal operation, that is to say with the cut-off frequency of 2 KHz.Such relationships can occur in particular at the start of the basebandchain where, on the one hand, AC coupling is required, since thefrequency converters for conversion of the radio frequency to basebandhave manufacturing-dependent DC offsets while, on the other hand,adjacent channels still exist without being filtered, since the ACcoupling is carried out directly at the output of the mixers, and thuseven before the channel filter in baseband. FIG. 4 describes this signalvoltage as a function of the time at the output of the baseband chain,that is to say downstream from the programmable amplifier and upstreamof the analog/digital converters.

FIG. 5 shows a filter arrangement which has been developed from thatshown in FIG. 1 and which, in addition to the high-pass filter R1, R2,C1, C2 with a switchable cut-off frequency, has a low-pass filter R3,C1; R4, C2 which can be connected and is connected upstream of thishigh-pass filter. In this case, a series resistor is connected upstreamof each of the series capacitors C1, C2 in the balanced signal path, andis annotated R3, R4. A second switch S3, S4 is connected in parallelwith these series resistors R3, R4, respectively. The switch positionsof the switches S1 to S4 are shown during normal operation in thepresent FIG. 5.

During a changeover mode, firstly, as already explained with referenceto FIG. 1, the higher cut-off frequency is activated for the high-passfilter by closing the switches 51, S2, while the low-pass filter formedby R3 with C1 and R4 with C2 is additionally activated by opening theswitches S3, S4. The precise switching sequence must in this case becarried out in order to avoid the sampling effect with the controlcircuit which drives the switches S1 to S4, such that the switches, S3,S4 are opened first of all, after which the switches S1, S2 are closed,for switching from normal operation to the changeover mode. Whenreturning from the changeover mode to normal operation, the switches S1,S2 are opened again first of all, and the switch S3 and the switch S4are closed only after this has occurred, in order to avoid the samplingeffect. The charge on the capacitors C1, C2 is accordingly changedduring the changeover mode with the time constants formed from theproduct of the resistance and capacitance, R3*C1 or R4*C2. This ensuresthat no sampling effect occurs at the output of the filter arrangement,even if the useful signal is not significantly less than the sudden DCchange caused by switching the gain.

FIG. 6 shows the signal profile D at the output of a baseband chain whenusing a circuit as shown in FIG. 5. This shows that the sampling effectno longer occurs. The stabilization processes that can still be seenduring the changeover mode are caused by the channel selection filterand/or the baseband filter. These can be reduced considerably, forexample, by reducing the gain after the channel selection, that is tosay by means of the conventional programmable amplifier there, or can beblanked out by insertion of a further AC coupling in the signalprocessing chain.

FIG. 7 uses an example of a simplified block diagram to shown a homodynereceiver arrangement with AC coupling which is, for example, in the formof a filter arrangement as shown in FIG. 1 or as shown in FIG. 5. Thereceiver is in this case in the form of an IQ receiver for processing ofcomplex-value signals and, in addition, for carrying the in-phase andquadrature components in each case as balanced signals.

In detail, an IQ mixer is provided for frequency conversion 1, and canbe supplied with a received radio-frequency signal, in each case in theform of a balanced signal, at in each case one signal input. The mixers1 also each have a local oscillator input, to which a local oscillatorsignal is supplied on the one hand unchanged and on the other hand witha phase shift of 90.degree. A baseband signal is produced on the outputside of the frequency converters 1, and is likewise in the form ofbalanced signal. A low-noise variable gain baseband amplifier 2 isconnected to the outputs of each of the frequency converters 1 in thein-phase and quadrature paths of the receiver. A high-pass filter with avariable cut-off frequency for AC coupling is connected on the outputside to the low-noise baseband amplifier 2. A baseband filter 4 iscoupled to each of the outputs of the high-pass filters 3 for channelselection, and each output of said baseband filter is in turn connectedto a programmable amplifier or PGC, Programmable Gain Control 5. On theoutput side of the programmable amplifiers, that is say at the output ofthe baseband signal processing chain that has been explained, a filteredand amplified baseband signal is produced as a complex-value signal,broken down into an in-phase component I and a quadrature component Q,with the I and Q components each being in the form of balanced signals.

The low-noise baseband amplifiers 2, the filter arrangements 3 and theprogrammable amplifiers 5 each have a control input which is connectedto a control circuit 6. The signal for controlling the gain of thelow-noise baseband buffer 2 is annotated E, the signal for varying thedesired cut-off frequency of the high-pass filter 3 is annotated F, andthe signal for varying the gain of the programmable amplifier 5, inparticular the minimum gain which can be set for it, is annotated G.

The method of operation of the circuit illustrated in FIG. 7, which issuitable for receiving signals coded using the code division multipleaccess method and which accordingly can be used for UMTS receivers, isevident from the signal profiles of the control signals E, F, G shown inFIG. 8.

FIG. 8 shows the signal profile of the control signal E for thelow-noise baseband amplifier 2 which, at the time, T0, is reducing thegain by 6 dB owing to a change in the reception conditions, for example,a greater received field strength. The −6 dB step in this case has asoft gain transition. At the same time as the change to the gain, thefilter arrangement 3 is switched to a higher cut-off frequency for avariable time interval .DELTA.T, that is to say the high-pass filters inthe filter arrangements 3 change from normal operation to the changeovermode, starting at the time T.sub.0 and for the time period .DELTA.T. Ifthe filter arrangement as shown in FIG. 3 is additionally designed witha low-pass filter characteristic as shown in FIG. 5, then this low-passfilter is connected upstream of the high-pass filter for the time period.DELTA.T. The programmable amplifier 5 at the end of the baseband signalprocessing chain is likewise switched by the control signal G to theminimum gain which can be set for it for the time period .DELTA.T andstarting at the time T.sub.0 in order for it not to amplify any furthera switching transient, which is invariably unavoidable even if it isvery small.

The combination of the accelerated AC coupling with the higher cut-offfrequency, the soft switching of the gain as shown by the signal E andthe switching off of the downstream amplifier stage 5 for thestabilization time .DELTA.T which is now short of the acceleratedhigh-pass filter as shown by the control signal G results in thedescribed advantages, that is to say a low cut-off frequency duringnormal operation of the high-pass filter 3, a short stabilization timeafter a change to the gain, negligibly small transients relating to theswitching times for the gain of the amplifier 2, an optimum drivecapability for downstream analog/digital converters with the usefulsignal, and thus an implementation with a small number of bits as wellas the capability to switch the gain of the amplifier 2 during normalreception.

Instead of the described process of switching the gain of the downstreamamplifiers 5 back in accordance with the control signal G, further ACcouplings may also be inserted into the baseband chain in alternativeimplementations.

What is claimed is:
 1. A receiver arrangement with AC coupling,comprising: an input configured to receive a radio-frequency signal; afrequency converter coupled to the input and configured to produce abaseband signal at its output; a variable gain baseband amplifiercoupled to the output of the frequency converter, and having a controlinput for varying a gain thereof; a filter arrangement for AC coupling,comprising an input coupled to an output of the variable gain basebandamplifier, the filter arrangement comprising: a high-pass filterconfigured to filter the baseband signal, having a cut-off frequencythat is switched between at least two values, with a lower of the atleast two cut-off frequencies used during normal operation and with ahigher of the at least two cut-off frequencies used in a changeovermode, and having a control input for varying the cut-off frequency; anda selectively activatable low-pass filter having a control input; acontrol circuit coupled to the control inputs of the variable gainbaseband amplifier and the filter arrangement, and configured toactivate the higher cut-off frequency of the high-pass filter andactivate the low-pass filter during the changeover mode based on areadjustment of the gain of the baseband amplifier.
 2. The receiverarrangement of claim 1, wherein the control circuit is configured toactivate the higher cut-off frequency in the high-pass filter of thefilter arrangement during a variable time interval, wherein the variabletime interval starts at a time of a gain change of the basebandamplifier based on the readjustment of the gain.
 3. The receiverarrangement of claim 1, further comprising a programmable amplifiercoupled to the output of the filter arrangement and to the controlcircuit, wherein the control circuit is configured to set a lowest gainthat can be set for the programmable amplifier during the changeovermode.
 4. The receiver arrangement of claim 3, further comprising abaseband filter configured to perform channel selection, wherein thebaseband filter is coupled between the filter arrangement and theprogrammable amplifier.
 5. The receiver arrangement of claim 1, whereinthe receiver arrangement is configured to process balanced signals. 6.The receiver arrangement of claim 1, wherein the high-pass filter in thefilter arrangement comprises a balanced signal path having therein aparallel branch that includes a first resistor and a parallel-connectedfirst switch, the balanced signal path further including a series branchconnected upstream of the parallel branch, wherein the series branchincludes a first capacitance.
 7. The receiver arrangement of claim 6,wherein the series branch of the balanced signal path in the filterarrangement further comprises a first parallel circuit comprising asecond resistor and a parallel-connected second switch that forms withthe first capacitance a portion of the selectively activatable low-passfilter, and wherein the parallel circuit is coupled upstream of thefirst capacitance.
 8. The receiver arrangement of claim 7, wherein thecontrol circuit is configured to open the parallel-connected secondswitch and close the parallel-connected first switch in the changeovermode, and open the parallel-connected first switch and close theparallel-connected second switch in the normal mode.
 9. The receiverarrangement of claim 7, wherein the parallel branch further comprises athird resistor and a parallel-connected third switch, connected inseries with the first resistor and the parallel-connected first switch.10. The receiver arrangement of claim 9, wherein the filter arrangementfurther comprises a second parallel circuit comprising a fourth resistorand a parallel-connected fourth switch that forms with a secondcapacitance another portion of the selectively activatable low-passfilter, and wherein the control circuit is configured to open theparallel-connected second switch and the parallel-connected fourthswitch and close the parallel-connected first switch and theparallel-connected third switch in the changeover mode, and open theparallel-connected first switch and parallel-connected third switch andclose the parallel-connected second switch and the parallel-connectedfourth switch in the normal mode.
 11. The receiver arrangement of claim1, wherein the higher cut-off frequency is greater than or equal toabout 1 Megahertz.